Feature: Electronic design
An undesirably high output impedance is a significant
disadvantage of all forms of the cascode transimpedance stage. Tis is worsened if two-pole minor negative feedback loop compensation is used. Two-pole minor loop compensation applies far less minor loop gain around the TIS across the audio band than single-pole minor loop compensation, and this causes the cascode TIS’s high output impedance at infrasonic frequencies to be sustained across most of the audio band (Figure 11). If two-pole minor negative feedback loop compensation is used
with the cascode TIS of Figure 10, then using a Class A MOSFET source follower to completely isolate the cascode TIS from the non-linear loading of the output stage may be unavoidable. Tis is because the output impedance of the cascode TIS of Figure 10 is very high, being, for example, roughly 1.4MΩ at 400Hz, which requires a buffer with an input impedance of at least 14MΩ to minimise the loading on the TIS. Regrettably, using an emitter-follower to bootstrap a cascode
TIS’s resistive output load so as to buffer it from the non-linear input impedance of a Class B output stage, as implemented in the compound emitter-follower/common-emitter TIS of Figure 1, does not sufficiently mitigate the problem. Tis is because the output impedance of the bootstrapping emitter-follower remains unacceptably high at low audio frequencies with single-pole minor negative feedback loop compensation and across most of the audio band with two-pole minor negative feedback loop compensation. Tis is not the case with the circuit of Figure 1 where, in contrast, the output impedance of the bootstrapping emitter-follower Q8 is small enough across the audio band to typically render negligible the non-linear loading of, for example, a Class-B double emitter- follower output stage upon emitter-follower Q8.
Stabilisation Without the extra compensation elements to be described here, the minor feedback loop of the circuit in Figure 10 has a negative phase margin and is, therefore, unstable. To stabilise the minor loop, the TIS is degenerated with resistor R14, which reduces minor loop gain, while emitter-coupling capacitor C1 introduces a leſt half plane (LHP) zero near the unity minor loop gain frequency, further enhancing stability margins. However, while increasing the stability margins of the minor
negative feedback loop, degeneration resistor R14 lowers the frequency of the right half plane (RHP) zero in the major negative feedback loop. Te RHP zero has the magnitude response of a LHP zero and the phase response of a LHP pole. In other words, at the frequency of the RHP zero, the major loop gain increases and is accompanied by a decrease in phase shiſt. Te RHP zero is undesirable because it may reduce stability margins significantly if it is sufficiently close to the unity major negative feedback loop- gain frequency. Te RHP zero arises because of the feed-forward path provided by Miller (minor loop) compensation capacitor C2 from the input to the output of the TIS. Te RHP zero may be eliminated by blocking the feed-forward
path from input to output of the TIS. Tis may be done by connecting C2 to the output of a unity voltage gain buffer whose
input is connected to the output of the TIS. Tis buffer can simply be a Class-A-MOSFET source follower connected between the output of the TIS and the input to the output stage transistors, but this defeats the object of using a push-pull TIS to drive the Miller compensation capacitor, as it is then driven by the output of the source follower and not the output of the TIS. Rather than eliminate the RHP zero by buffering C2 from the
output of the TIS, R23 was instead connected in series with C2 and made sufficiently large so as to cause the zero to migrate from the RHP to the LHP, thereby improving the stability margins of the major negative feedback loop. Unfortunately, R23 rather reduces the stability margins of the minor negative feedback loop. Tese stability margins are further improved by introducing
feed-forward capacitor C6 to bypass the four transistors in the non-inverting signal path of the differential TIS at high frequencies. Te feed-forward path defined by C6 introduces a LHP zero in the vicinity of the unity minor loop gain frequency, and, consequently, emitter-coupling capacitor C1 may not be required aſter all. Alternatively, C6 may be used to bypass just the first three of the
four transistors in the non-inverting signal path of the differential TIS at high frequencies, by connecting it to the emitter of the last common-base transistor Q18. Tis has the advantage that C6 has only Q18’s base-emitter voltage across it, while connecting it to the input of the TIS’s current mirror as shown in Figure 10 means it has close to the entire negative DC supply voltage across it. Either approach may be adopted, with both giving virtually identical improvements in minor loop stability margins. Note that the RHP zero caused by Miller compensation in the
preferred topology of Figure 1 can usually be ignored, provided the TIS’s quiescent current is sufficiently large – on the order of milliamps, because it is typically located at a frequency more than four decades higher than the unity major loop gain frequency.
A novel circuit Incidentally, the feed-forward compensation technique described here may also be used to dramatically improve the stability margins of the compensating minor negative feedback loop in the novel differential boosted-current complementary folded cascode TIS of Figure 13. In this arrangement, the two feed-forward capacitors C2 and C3 bypass the TIS’s complementary differential pairs at high frequencies, driving its current mirrors directly, and thereby introduce a beneficial LHP zero in the vicinity of the unity minor loop gain frequency. Te TIS of Figure 13 is twice as efficient as that of Figure 10,
because the maximum peak output current available from the TIS of Figure 13 is equal to twice the quiescent current of each of its tail current sources Q38 and Q39, assuming that both provide equal amounts of current. On the other hand, the maximum peak output current available from the TIS of Figure 10 cannot exceed the value of its tail’s quiescent current. Two current sinks instead of one constitute the tail of the TIS in Figure 10 because this halves the power dissipated in each current sink. Note that current sources Q12 and Q13, which provide
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